Solid state chopper ballast for gaseous discharge lamps

ABSTRACT

A single phase, high frequency, transistor or gate turnoff thyristor chopper ballast circuit especially suited for mercury vapor lamps preferably operates on a unfiltered full wave rectified line voltage and electronically shapes the lamp current and therefore the line current to obtain a high power factor. The ballast circuit is lightweight with low volume due to elimination of large low frequency energy storage, filtering, and transformer components. The forced, high frequency ripple lamp current waveshape, achieved by comparison of the sensed current with an appropriate reference signal, provides for good regulation, an initially high starting current to eliminate glow-to-arc mode, automatic sweeping of the chopping frequency to avoid acoustic resonance effects, and a minimum current in the valley regions of the supply voltage for improved reignition characteristics.

United States Patent Park et al.

[ June 17, 1975 SOLID STATE CHOPPER BALLAST FOR GASEOUS DISCHARGE LAMPSInventors: John N. Park, Rexford; Steven C.

Peak, Schenectady; Robert L. Steigerwald, Scotia, all of NY.

General Electric Company,

Schenectady, N.Y.

Jan. 2, 1974 US. Cl 315/208; 315/209 R; 315/224;

Int. Cl "b 41/24; GOSf 1/08 Field of Search 315/200 R, 207, 208, 224,

315/287, 283, 307, D16. 5, DIG. 7, 209 R References Cited UNITED STATESPATENTS [73] Assignee:

[22] Filed:

[21] Appl. No.: 430,088

Primary Examiner-Palmer C. Demeo Assistant ExaminerE. R. LaRocheAttorney, Agent, or FirmDona1d R. Campbell; Joseph T. Cohen; Jerome C.Squillaro [57] ABSTRACT A single phase, high frequency, transistor orgate turnoff thyristor chopper ballast circuit especially suited formercury vapor lamps preferably operates on a unfiltered full waverectified line voltage and electronically shapes the lamp current andtherefore the line current to obtain a high power factor. The ballastcircuit is lightweight with low volume due to elimination of large lowfrequency energy storage, filtering, and transformer components. Theforced, high frequency ripple lamp current waveshape, achieved bycomparison of the sensed current with an appropriate reference signal,provides for good regulation, an initially high starting current toeliminate glow-to-arc mode, automatic sweeping of the chopping frequencyto Powell, Jr. 315/207 X avoid acoustic resonance effects, and a minimumcur- Gc 315/233 X rent in the valley regions of the supply voltage forim- Mahler 315/D1G. 5 proved reignifion characteristics- Rosa 3l5/D1G. 5

15 Claims, 9 Drawing Figures H/Gl-fiff/VCY 32 IPIFZWZ/VCI (WM wasHut/M)? SOLID STATE CHOPPER BALLAST FOR GASEOUS DISCHARGE LAMPSBACKGROUND OF THE INVENTION This invention relates to a solid stateballast circuit for gaseous discharge lamps, and more particularly to ahigh frequency chopper ballast for mercury vapor lamps which utilizeselectronic techniques to shape the line current for high power factorand to obtain good regulation.

The majority of mercury lamps presently in use employ electromagneticballasts with bulky low frequency transformers, inductors and largepower factor correcting capacitors. Although a number of circuits usingsolid state devices have been developed for ballasting high intensitydischarge mercury lamps and similar lamps, those circuits which operateon 60 Hz alternating-current or full wave rectified voltage incorporatebulky and expensive components. More sophisticated high frequencycircuit approaches do not achieve an economic solution to the problemand ignore some of the major problem areas such as acoustic resonanceeffects and electrode degradation clue to arc initiation.

The combination of features an electronic ballast desirably should haveare to provide high power factor, high efficiency, low acoustic andradio frequency interference noise, and good regulation in a singlephase circuit without requiring heavy power frequency magnetics andlarge correction and energy storage capacitors. Further, the ballastcircuit should be relatively insensitive to normal line transients, thelamp should not extinguish upon rapid excursion to 65 percent of ratedline voltage, lamp operation should avoid visible flicker or acousticresonance effects caused by continuous operation at a constant highfrequency, and circuit operation should be stable for very long periodsof time. The circuit should operate over an ambient temperature range of30C to +85C and provide negligible electrical interference to itssurroundings.

In the concurrently filed application Ser. No. 429,9l4 by Robert L.Steigerwald and John N. Park, entitled Power Circuits for Obtaining aHigh Power Factor Electronically," and assigned to the same assignee, anumber of single phase chopper circuits for alternating-current anddirect-current loads are described which use only high frequencyfiltering and electronically shape the line current to obtain a highpower factor. As a typical application, a mercury lamp ballast circuithaving many of the foregoing desirable features is disclosed. Thepresent application relates to an improvement on this ballast circuitwith emphasis on obtaining good lamp operation in a more satisfactorycircuit configuration.

SUMMARY OF THE INVENTION The new solid state, high frequency chopperballast is suitable for energization by unfiltered low frequencyalternating-current line voltage, preferably full wave rectified withonly high frequency filtering, and broadly includes a controlledswitching means, such as a power transistor, and coasting device means,such as a power diode, that conduct alternately and supply lamp currentthrough a coasting inductor to a mercury vapor lamp or other gaseousdischarge lamp. A current sensor is coupled to sense the instantaneous,high frequency ripple lamp current. The control circuit has provisionfor generating a preselected reference signal waveshape to determine thepower level, optionally regulate the lamp current, and to effect shapingof the lamp current and therefore the line current to obtain a highpower factor. I By effectively comparing the sensor and referencesignals, an output signal is produced for controlling the application ofalternate turn-on and turnoff signals to operate the controlledswitching means at a variable high frequency chopping rate to shape thelamp current as determined by the reference signal waveshape. As aresult of automatic sweeping of the chopping frequency and as a resultof the low ripple amplitude, acoustic resonance effects are avoided.

In accordance with the invention, an improved lamp current waveshape isobtained at initial start-up of the lamp. Also, the lamp current isimproved by supplying in a more satisfactory manner a minimum lampcurrent in each cycle when the comparing means is ineffective to shapethe lamp current, i.e., during the valleys or low voltage regions of thepulsating or sinusoidal power voltage. To avoid the undesirableglow-to-arc mode, the control circuit has provision for temporarilyshaping the reference signal at initial start-up to obtain a highstarting current, as by using a long time constant network to modify theaction of the control function generator in the reference signalgenerating means. Minimum lamp current in the valley or low voltageregions for improved reignition characteristics is supplied by the highfrequency filter and, in the preferred transistor d-c chopper ballast,by using local energy storage capacitors in the improved transistordrive curcuit power supply to provide base current to the normalconducting positive base drive circuit to maintain power transistorconductivity. Other control circuit improvements include a low voltagepower supply for the comparator which supplies clipped, regulatedvoltage except during the valley regions when it is not needed, therebyeliminating the need for electrolytic capacitors. An improved transistorbase drive circuit and power supply therefor are also disclosed. The newhigh frequency chopper ballast for mercury lamps incorporates thedesirable features previously mentioned, is highly efficient with lowvolume and light weight, and does not employ low frequency energystorage and correction capacitors, inductors, and power transformers.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a simplified schematiccircuit diagram partly in block diagram form of a d-c chopper ballastfor a mercury vapor lamp and is used to explain the principles of theinvention;

FIG. 2 is a waveform diagram of a sinusoidal reference signal withclosely adjacent control band limits for controlling the intervals ofconduction and nonconduction of the power transistor in FIG. 1;

FIG. 3 is a schematic power circuit diagram with control circuitconnections according to the preferred embodiment of the mercury lampsolid state ballast circuit;

FIG. 4 shows ideal waveform diagrams of the line current and voltage,lamp current, and reference signal for the preferred ballast circuit;

FIG. 5 is an enlarged diagram of the flattened sinusoidal referencesignal and control band limits for the control circuit logic signals;

FIG. 6 are typical oscilloscope waveforms of the lamp voltage and lampcurrent illustrating, at an enlarged scale, the high frequency rippleproduced by operation of the chopper ballast;

FIG. 7 is a detailed control circuit schematic diagram for the mercurylamp ballast circuit;

FIG. 8 is a diagrammatic side view of a transformer with a pair ofsecondary windings for supplying power to the logic and power transistorbase drive circuits in FIG. 7; and

FIG. 9 is a sketch of a portion of the control function generator inFIG. 7 modified to obtain auxiliary adaptive control of the mercury lampchopper ballast, for example, in response to sensing the ambient lightlevel.

DESCRIPTION OF THE PREFERRED EMBODIMENT The high frequency, singlephase, direct current chopper circuit shown in FIG. I supplies acontrolled current waveshape and controlled power to a mercury vaporlamp or other appropriate gaseous discharge lamp, and the line currentis accordingly electronically shaped to obtain a high power factor. Thepower circuit is relatively simple and economical, and uses no bulkysupply frequency transformers, inductors, or large energy storage orpower factor correcting capacitors. The control circuit operates on thebasis of continuously comparing the sensed lamp current with apreselected reference signal waveshape to thereby determine the highfrequency switching rate of the power transistor and generate thedesired lamp current waveshape. In the preferred chopper ballast ofFIGS. 3-9, other desirable operating characteristics such as goodregulation, a good starting current waveform, etc., are provided as willbe explained.

The single phase power circuit (FIG. 1) has a pair of input terminals 20and 21 connected, by way of illustration, to a 60 Hz, 277 volt source ofalternating current, but other power frequencies and voltages can beused depending on the application. A diode bridge rectifier 22 connectedto the a-c input terminals produces a full wave rectified sinusoidalvoltage which is supplied essentially unfiltered to the chopper circuit.A high frequency filter provided for example by a series inductor 23 anda shunt capacitor 24 is connected across the output terminals of thebridge rectifier 22, but these high frequency filter componentsessentially are provided to isolate the high frequency chopping from the60 Hz line. It may be preferable to further include a second shuntfilter capacitor connected between the input lines, and other variationsare possible depending upon the amount of line filtering required. Inthe chopper circuit, a power transistor 25 and power coasting diode 26are connected in series between the high voltage, 120 Hz, pulsating d-csupply terminals 27 and 28, and a coasting inductor 29 is connected inseries with the mercury lamp 30 across the coasting diode 26. A suitableload current sensor 31, such as a small current transformer or sensingresistor, is coupled in series with the lamp 30, and continuouslysupplies an input signal to the control circuit which is indicative ofthe magnitude of the instantaneous lamp current. In operation, in thesame manner as a time ratio control circuit, the power transistor 25 isturned on and off at a high frequency switching rate. During conductingintervals of the transistor 25 power is supplied to the load 30 throughthe coasting inductor 29, and during nonconducting intervals of thetransistor 25 the coasting diode 26 becomes forward biased and providesa path for load current as the stored energy in coasting inductor 29discharges. The circuit is preferably operated in the tens of kilohertzfrequency range, in the range of about 10 kHz to 40 kHz for thisapplication. With this power circuit configuration, it is noted, thereis inherently a small high frequency ripple in the load current.

The coasting diode 26 and power transistor 25 are preferably matcheddevices in order to eliminate additional power circuit components in thecoasting path. In each high frequency cycle when the power transistor isrendered conductive, the coasting diode does not immediately block dueto stored charges and higher than normal currents flow in the powertransistor. The peak current generated during this transient is limitedby employing a fast recovery coasting diode and by making a reasonablyclose match of the turn-on time of transistor 25 to the recovery time ofthe coasting diode. A controlled recovery diode is used rather than asnap off diode to prevent large transient voltages from developingacross the diode and to prevent generation of high frequencydisturbances.

The control circuit generates a reference signal which is basically inphase with the applied line voltage and has a predetermined waveshapeand magnitude to achieve high power factor and deliver a selected amountof power to the load. As has been pointed out, in this power circuit thereference signal determines the load current waveshape and thus the linecurrent waveshape and input power for a given lamp. The reference signalwaveshape can also be selected to achieve additional desirable featuressuch as good regulation and suitable load current waveshapes to meet therange of load operation conditions. Accordingly, the exact referencesignal waveshape that is selected depends upon the combination offeatures that are required or the best compromise, depending upon theparticular circumstances. In order to eliminate the need for specialsignal generating equipment such as low frequency oscillators, thecontrol signal is derived directly from the a-c input lines and thenshaped according to a selected control function to obtain the desiredreference signal waveshape. The reference signal is then also in phasewith the line voltage. To this end, a step-down transformer 32 isconnected across the input lines and, for the power circuitconfiguration, feeds a diode bridge rectifier 33 so that the input to acontrol function generator 34 is a full wave rectified d-c voltage.Generally speaking, the control function is selected as previouslydescribed and can be a constant gain, an electronically variable gain, asquaring circuit, a square root circuit, etc., depending upon the typeof load and control desired. Referring also to FIG. 2, there are closelyadjacent control band limits associated with the reference signal thatefi'ectively determine the limits of excursion of the lamp current asshaped by the controlled switching action of the power transistor 25.The control band is effectively placed about the reference signal, orcan be entirely at one side of the reference signal or closely spacedfrom it. In any case, the control band limits are close to or coincidewith the reference signal and conform to its waveshape. Although othercircuitry can be employed to obtain the control band limits, a simpleand effective implementation is by the use of a comparator 35 withhysteresis. The hysteresis characteristic may be obtained by a feedbackconnection from the output of the comparator to the positive input ofthe comparator, as is further explained with regard to FIG.

7. The reference signal is applied to the positive input of comparator35, while the negative input is a sensor signal indicative of theinstantaneous lamp current generated by the current sensor 31.

An output from the comparator 35 is amplified by amplifier 36 and iseffective to apply a base drive signal to the power transistor 25 todrive it into saturation and render it conductive. Assuming that lampcurrent is circulating in the coasting path and is decreasing, and thatthere is a low output from the comparator 35 so that power transistor 25is turned off, the lamp current continues to decrease until the currentsensor signal at the negative input of the comparator is equal to andabout to go below the reference signal control band limit at thepositive input of the comparator (i.e., the reference signal minushysteresis). A comparator output is now produced, turning on the powertransistor 25 and causing an increase in the lamp current as current isdrawn from the supply. The reference signal now switches to its uppercontrol band limit value (i.e., the reference signal plus hysteresis),and the comparator output remains high and the power transistor 25remains conductive until the lamp current increases and the currentsensor signal becomes equal to the value of the other reference signalcontrol band limit. The comparator output then goes low, thereby turningoff the power transistor 25 and switching the value of the referencesignal at the positive input of the comparator to its lower control bandlimit. The lamp current therefore has a small amount of ripple about thenominal value determined by the reference signal hysteresis. Thechopping frequency of the circuit is not constant during each cycle ofthe rectified sinusoidal voltage supplied to the chopper circuit. Thechopping frequency is determined primarily by the value of the coastinginductor 29, the instantaneous voltage difference between the rectifiedsinusoidal voltage feeding the chopper and the actual lamp voltage, thestorage time of power transistor 25, and the comparator hysteresis. Forthe circuit shown in FIG. 1, the chopping frequency is considerablyhigher at the middle of the half cycle than at either end where thesupply voltage is low. This periodically variable chopping frequency isdesirable for some loads, for example as a factor in eliminatingacoustic resonance problems in mercury vapor lamps which can occur undercertain constant high frequency conditions.

Having discussed the underlying principles of the chopper circuit andcontrol technique, the preferred embodiment of the invention will bedescribed with regard to FIGS. 3-9. The power circuit (FIG. 3) issimilar to FIG. 1 and is suitable for HID mercury lamps in the 74 to1000 watt range and also, without modification, for other gaseousdischarge lamps that are operable on unfiltered full wave rectified 120Hz (or 100 Hz) voltage which cyclically drops to zero in the valleyregions. The preferred circuit is discussed with regard to ballasting a250 watt mercury vapor lamp, giving typical values of the voltages,currents and other parameters to clarify the presentation. Whenappropriately modified, the ballast circuit may be used with stilldifferent types of gaseous discharge lamps that require a supply of lampcurrent in the valleys to maintain sufficient lamp ionization until the120 Hz wave rises to a usable level. Within the broader scope of theinvention, the chopper ballast can be constructed in a-c versionswithout a full wave rectifier using a pair of inverse-parallel powerswitches and coasting devices to provide bidirectional conductingcapability. This is further explained in the aforementioned concurrentlyfiled application Ser. No. 429,914, to which the reader may refer forfurther information. lnstead of the power transistor, a gate turnoffthyristor can be used, both of these being described generically as acontrolled solid state switch with a single electrode for turn-on andturnoff. The power circuit is preferably fabricated by power moduletechniques while the control curcuit is fabricated using integratedcircuit and microelectronic techniques.

In FIG. 3, high frequency filtering and transient voltage protection isprovided at the input to the chopper circuit and is effective as to boththe power circuit and the control circuit. The input high frequencyfilter serves primarily to limit the amount of radio frequencyinterference which appears across the line due to the operation of thechopper, and includes a second shunt capacitor 24' as well as the seriesfilter inductor 23 which now has a small parallel resistor 40 to preventringing in the filter circuit due to transient excitation. The filterinductor also provides sufficient series impedance to permit effectiveline transient voltage sup pression for all the power and controlcircuit components by means of a single polycrystalline varistor 41effectively connected between the input terminals of diode bridgerectifier 22. The input voltage for the control circuit 42 is takenbetween these same two lines. By way of example, varistor 4] is a GE-MOVvaristor (trademark of the General Electric Company), type V275LA20. Thehigh frequency filter also includes the shunt capacitor 24 (now providedwith a parallel bleeder resistor 43) to provide a circulating path forthe high frequency current components of the transistor chopper circuit.Thus, the voltage feeding the chopper is essentially a full waverectified Hz line voltage.

The power transistor 25, for instance, is a Toshiba ZSCl 172A transistorand a suitable matched coasting diode 26 is a MR856 power diodemanufactured by Motorola, lnc. The current sensor is a small sensingresistor 31', such as a one-half ohm resistor, connected in series withcoasting inductor 29 and mercury lamp 30, the coasting diode 26 beingconnected across all these elements. The voltage across the sensingresistor 31 is supplied to the control circuit 42 and is indicative ofthe instantaneous lamp current. This is a negativegoing signal voltagein this circuit arrangement. The input voltage derived from the linesupplies power to the control circuit 42 and also provides a controlsignal that is modified by the selected control function to provide thereference signals. The control circuit 42 further includes dual basedrive circuitry for the power transistor 25 which is effective to turnon, hold on, positively turn off, and hold off the power transistor. Thebase drive current and voltage supplied by control circuit 42 providesthe proper conditions for chopper operation with a full wave rectifiedsupply voltage. As will be further explained, the base current isproportional to the collector current in the power transistor, andelectrolytic capacitors are not needed in the base drive circuitry, noralso in the control function generator and comparator circuitry.

Referring to the waveform diagrams in FIG. 4, it is realized in practicethat the line voltage varies under normal conditions. The referencesignal e is a full wave rectified, flattened sinusoidal signal, and thecontrol function additionally provides an electronically variable gaincharacteristic so that the lamp current remains approximately constantdespite line voltage variations. This provides good lamp currentregulation for a reasonable range of line voltage variations. A mercurylamp load is a non-linear load with a negative resistance characteristicat low frequencies, and further has some of the characteristics of aback emf load. There is some lamp current at the beginning of each cyclebefore ignition and at the end of each cycle, when the line voltage islow. To further explain the concept of the back emf load, if it isassumed that the load is a battery being charged, it is readily seenthat power is transferred to the battery only in those portions of thecycle when the instantaneous applied voltage is greater than the batteryvoltage. For instance, for a battery of I volts and a peak full waverectified sinusoidal voltage of 400 volts, no power is transferred tothe battery at the beginning and end of the cycle when the instantaneousvoltage is below I00 volts. The lamp voltage between the terminals ofordinary mercury vapor lamps is typically about I30 volts. It will befurther understood that there is an impedance transformation by virtueof the operation of the chopper circuit, so that the lamp current andthe line current do not necessarily have the same waveshape ormagnitude. From the foregoing example, it is seen that there is avoltage transformation, and in like manner, there is also a currenttransformation. Based on the foregoing analysis, there is some lampcurrent at the beginning and end of each cycle when the supply voltageis low, and in the inter mediate portion of each cycle the lamp currentis forced to follow the flattened sinusoidal reference signal. Theshaped line current draws increased current due to ignition of the lampnear the beginning of the cycle, but can be described as being roughlyconstant in the intermediate portion of each cycle, dropping at the endof the cycle in the valley regions of the rectified supply voltage. Thisline current waveshape is in phase with the line voltage and provideshigh power factor, easily in excess of 90 percent, with good regulationof the lamp current and input power.

The flattened and regulated sinusoidal reference signal is actuallynegative-going as shown in FIG. 5. The lamp current has a high frequencyripple about a nominal value, and in the reproductions of typicaloscilloscope waveforms of the lamp current given in FIG. 6, the ripplein the shaped, flattened sinusoidal lamp current is illustrateddiagrammatically at enlarged scale. The lamp voltage waveform alsoexhibits a high frequency ripple and shows the momentarily highervoltage drawn at reignition at the beginning of each cycle. In thevalleys of the full wave rectified supply voltage, the lamp plasmaactually deionizes to a certain extent, such that it can be said thatthe lamp reignites in each cycle. The minimum lamp current in thevalleys maintains sufficient ionization for good reignitioncharacteristics. For a 277 volt to 208 volt source, either 60 Hz or 50Hz, the supply voltage rises to a sufficiently high level near thebeginning of a cycle to permit reignition.

FIG. 7 is a detailed schematic circuit diagram of the improved controlcircuit 42. The step-down transformer 45 is energized by the highfrequency filtered, varistor-protected line voltage and has a pair ofcentertapped secondary windings, one of which supplies low voltage, highcurrent power (typically l2 volts peak, I amp peak) for the dual basedrive circuitry of power transistor 25, while the other pair ofcenter-tapped secondary windings supplies high voltage, low currentpower (typically volts peak, 30 milliamps peak) for the logic portionsof the control circuit. A suitable transformer construction thatprovides a low capacitance between the primary winding and eachsecondary winding is shown in FIG. 8. The bobbin 47 is disposed aboutthe central leg of the magnetic core 48. Bobbin 47 has a multiple wallstructure that provides a series of axially spaced compartments for thewinding of the separate transformer windings in the axial sequence ofS1, S3, P, S4, and S2. The secondary winding designations correspond tothose in FIG. 7. This low capacitance wafer wound design is effective toprevent the coupling of high frequency current components between thesecondary windings and between the secondary and primary windings, i.e.,it provides a low rfi coupling. The centertapped secondary windings S1and S2 (FIG. 7) are connected to a first full wave diode bridgerectifier 49 and generates a positive-going rectified sinusoidal voltageat one output junction 50 and a negative-going rectified sinusoidalvoltage at the other output junction 51. This negative rectifiedsinusoidal voltage, with a typical peak value of about 50 volts, is fedto the control function generator 34 which produces the flattenedsinusoidal, automatic gain controlled reference signal.

Control function generator 34 is comprised by a resistive voltagedivider connected between the junction 51 and a reference or common bus52 which includes the resistors 53-56. The generated reference signal istaken at the junction of resistors 55 and 56 and supplied to thepositive input of comparator 35. Flattening of the sinusoidal controlsignal is accomplished by a small resistor 57 and a small Zener diode 58connected in series between the junction of resistors 54 and 55 and thecommon bus. A small amount of current is diverted through this network.The automatic gain control feature is obtained by means of a MOS orinsulated-gate field effect transistor (FE'I') which is connected inseries with a potentiometer 60 across the resistor 56 and acts as avariable resistance in the shunt path. The peak voltage of thesinusoidal control voltage is detected by a peak detector circuit 61 anddetermines the gate voltage of FET 59. To this end, a high resistancevalue potentiometer 62 is connected between the junction of resistors 53and 54 and the common bus 52, and the voltage at the potentiometerpointer is supplied through a blocking diode 63 and a very largeresistor 64 to the peak detector 61, which is comprised by a largeresistor and a capacitor connected in parallel between the gate of PET59 and bus 52. In this arrangement, diode 63 prevents the capacitor fromdischarging rapidly. In operation, peak detector 61 changes the gatevoltage and hence the resistance of FET S9, and therefore the value ofthe shunt resistance path in the resistive voltage divider, so that thereference voltage at the junction of resistors 55 and 56 isapproximately constant despite variations in the peak value of thecontrol voltage due to line voltage variations.

An additional important function of the peak detector circuit 61 forcontrolling the gate voltage of FET 59 is to provide an improvedstarting current waveform for the lamp to minimize electrode degradationduring arc initiation. The time constant of the series RC network(primarily resistor 64 and the capacitor) is relatively long and iseffective to delay the divider action a few seconds. That is, thecapacitor at the gate of FET 59 charges slowly upon exciting the ballastcircuit, with the result that the FET resistance is initially high as isthe value of the generated reference voltage. The starting lamp currenttherefore is momentarily relatively high to quickly heat the cathode andavoid the undesirable glow-to-arc mode. By way of illustration, for a250 watt mercury lamp, the starting current ramps from amps peak to thenormal 3 amps peak in approximately 8 seconds. Also, it is possible toprovide the circuit with sophisticated adaptive control by controllingthe voltage at the gate of FET 59. For example, referring to FIG. 9, theambient light level may be sensed by a phototransistor 65 or otherphotosemiconductor and used to actuate an auxiliary adaptive controlcircuit 66 which in turn can determine the voltage at the gate of thefield effect transistor and consequently the magnitude of the referencevoltage. Alternatively, the output from the adaptive control circuit canbe connected directly to a comparator input. In this manner, a lamp canautomatically be turned on and off in response to the ambient lightlevel, or timing circuits may be employed to control the power level ofoperation over any given time duration, and there are otherpossibilities. Adaptive control circuit 66 can be employed to controlthe power delivered to the lamp. That is, by sensing the lamp voltage aswell as the lamp current, the sensed lamp voltage is used to control theoutput of auxiliary adaptive control circuit 66. The reference signal isthen modified by control of the gate voltage of PET 59 to keep the lamppower approximately constant.

The comparator 35 is preferably an integrated circuit component such asthe LM-3ll device manufactured by the National Semiconductor Corp. Apositive low voltage power supply and a negative low voltage powersupply produces the respective voltages +V,, and V,,, for supplyingpower to the comparator. These power supplies are produced by clippingthe high positive and negative voltages at the outputs of bridgerectifier 49, and obviate the need for electrolytic capacitors. Althoughthere is no power for comparator 35 during the valleys of the full waverectified 120 Hz voltage, power is not needed at these times since lampcurrent during the valley regions is not determined by action of thecomparator. Some lamp current is provided by another mechanism, as willbe explained. The positive low voltage power supply is comprised by aresistor 67 and a small Zener diode 68 conneceted in series between thebridge output junction 50 and the center-tap between the secondarytransformer windings SI and S2, and a transistor 69 having itscollector-base connected across the resistor 67 while the emitter isconnected to the appropriate pins in comparator 35. This is recognizedas being a series pass transistor regulator. As the voltage at point 5.rises, current is supplied through resistor 67 to Zener diode 68 and tothe baseemitter junction of transistor 69. After the Zener diode clampsthe base voltage, the circuit functions as a series voltage regulatorwith the voltage at the emitter of transistor 69 being approximately +5volts. The negative low voltage power supply at the other side of bridgerectifier 49 is similar with corresponding components designated bycorresponding primed numerals. In addition, it is noted that a highfrequency filter capacitor 70 is connected between the junction ofresistors 53'and 54 and the common bus 52 to filter undesirable highfrequency transients in the control voltage. Also, to provide filteringof the comparator power supply, small capacitors 71 and 71 arerespectively connected between the +V and buses and the common bus 52.

The minus terminal of sensing resistor 31 as previously mentioned, iscoupled to the negative input of comparator 35, while the plus terminalis referenced to the center-tap between the pair of secondary windingsS1 and S2 of transformer 45, which is the common point. To eliminatefast switching transients which could give a false peak lamp currentidentification, an RC filter comprised by resistor 72 and capacitor 73is effectively connected across the sensing resistor. The positive inputis connected through a resistor 74 to the junction of resistors 55 and56 at which the flattened sinusoidal reference signal is generated. Toprovide a comparator hysteresis characteristic, a relatively largeresistor 75 is connected in a feedback path between the output and thepositive input of the comparator, and functions with resistor 74 as avoltage divider. The amount of feedback voltage or feedback current atthe positive input has two values depending upon whether the comparatoroutput is high or low. The net instantaneous voltage at the positiveinput is thus determined by the instantaneous value of thenegative-going flattened sinusoidal reference signal and by the amountof feedback voltage. As a result of the normal operation of the choppercircuit, as previously explained, the changing current sensor signal inthe logic circuit at the negative input of the comparator is alternatelycompared with the two reference signal control band limits. This will bereviewed again later.

The output of comparator 35, which typically has a low output of 5 voltsand a high output of +5 volts, is coupled to an output transistor 76which provides the interface between the logic circuit and the powertransistor drive circuit. In particular, the comparator output isconnected to the junction of a pair of resistors 77 and 78 that areconnected between the base and an emitter resistor 93 for transistor 76,the resistor 93 further being connected to the V,,,. bus. Upon theoccurrence of a high comparator output, current is supplied totransistor 76 thereby rendering it conductive. The net effect of thisaction, it will be recalled, is to turn off the power transistor 25.

Relatively low voltage, high current, full wave rectified 120 Hzunidirectional voltage is supplied to the power transistor base drivecircuit by means of a sec- 0nd diode bridge rectifier 80 that isenergized by the second pair of secondary windings S3 and S4 oftransformer 45. A pair of relatively small, local energy storagecapacitors 81 and 82 are respectively coupled between the center-tap ofthe transformers secondary windings S3 and S4 and the positive andnegative d-c supply terminals 83 and 84 of bridge rectifier 80. In eachcycle, these capacitors store energy which is available for discharge inthe valley regions of the rectified 120 Hz wave, thereby providing asource of base current for the power transistor 25 in the valley regionswhen the control logic does not function. These capacitors also providelow impedance sources so that fast rising current waves can be developedto properly drive power transistor 25. The power transistor base drivecircuitry is divided into alternately operating positive and negativebase drive circuits 85 and 86 that serve to turn on, hold on, positivelyturn off, and hold off the power transistor 25. The magnitude of thepositive base current varies as a chopped half sinusoid since only highfrequency filtering is provided by the capacitors 81 and 82, andtherefore the collector current in power transistor 25 is proportionalto the base current in transistor 25. Thus, the peak base current (Iamp) is supplied only when it is absolutely needed at the point ofhighest collector current, while at other times base current is reduced,thereby obtaining high efficiency. in the positive base drive circuit85, the collectors of a pair of transistors in a Darlington amplifier 87are connected through a pair of parallel resistors 88 to the positivebridge output terminal 83, while the emitter of the Darlington amplifieris coupled to the base electrode 89 of the power transistor. The base ofthe first transistor is coupled through a biasing resistor 90 to theterminal 83, with the result that the transistor Darlington amplifier 87is normally conducting and supplies base current to the base electrode89. The negative base drive circuit includes a second Darlingtonamplifier 91 comprised by a pair of opposite type transistors whoseemitter and collector are respectively connected together and tied tothe base electrode 89. The emitter of the Darlington amplifier 91 iscoupled through a resistor 92 to the negative bridge supply terminal 84,and the base of the Darlington amplifier is coupled directly to thecollector of the transistor 76 and is also coupled directly to the baseof the other Darlington amplifier 87. With this arrangement, a positiveoutput from comparator 35 turns on the transistor 76, which is effectivein turn to turn off the Darlington amplifier 87 in the positive basedrive circuit while simultaneously rendering conductive the Darlingtonamplifier 91 in the negative base drive circuit. Excitation of thenegative base drive circuit 86, of course, renders the power transistor25 nonconductive. Upon application of the negative base drive current tothe base electrode, stored charge in the base of power transistor 25 isextracted and it turns off. During the remainder of the off-time,transistor 76 remains conductive since there is current through resistor90, and the small negative voltage at the junction of resistor 90 andthe collector of transistor 76, to which the base of Darlingtonamplifier 91 is connected, is effective to maintain the conductivity ofDarlington amplifier 91 and apply a negative bias to the base electrode89 which positively holds off the power transistor 25. With thisarrangement, clamping diodes between the base and emitter of the powertransistor 25 are not needed since base electrode 89 is effectivelyclamped to the V,,, supply through transistor 76 and resistor 93. In thevalley regions of the pulsating 120 Hz unidirectional voltage, the localenergy storage capacitor 81 discharges to provide a small amount ofcurrent through resistors 88 and 90 and Darlington amplifier 87 tomaintain the conductivity of power transistor 25 in the valley regions.The value of high frequency filter capacitor 24 in the power circuit asshown in FIG. 3 is sufficiently large (such as 3 microfarads for thecircuit being described) to maintain some lamp current in the valleys ofthe energizing power voltage, a condition which is desirable for goodreignition characteristics.

The operation of the solid state mercury lamp chopper ballast will bereviewed only briefly with reference to FIGS. 3-7. Since only highfrequency filtering ofthe line voltage is provided in the power circuit,the voltage supplied to the transistor chopper circuit is essentially apulsating, full wave rectified, 120 Hz sinusoidal volt age, The linevoltage is also supplied by means of stepdown transformer 45 to thecontrol circuit 42. In the control circuit (FIG. 7), the negative fullwave rectified, relatively high voltage, low current (50 volts, 30milliamperes) sinusoidal voltage at the output junction 51 of bridgerectifier 49 is used as a control voltage for the control functiongenerator 34. In this sub-circuit. a voltage divider comprised byresistors 53-56 has a variable resistance component provided by FET 59,the channel of which is connected in series with variable resistor 60across the resistor 56. The automatic gain control feature is obtainedsince the gate voltage as determined by the peak detector 61 isproportional to the peak of the rectified control voltage. When themagnitude of this voltage drops, for example, FET 59 tends to turn offand increases the variable resistance in the voltage divider so that thereference signal taken at the junction between resistors 55 and 56remains approximately constant with line voltage variations. Thesinusoidal control voltage is further flattened somewhat by means of asmall current drain through the resistor 57 and Zener diode 58. Theregulated, flattened sinusoidal reference signal supplied to thepositive input of comparator 35 results in good lamp current regulationand a slight reduction in the peak current which the power transistor 25conducts (as compared to the unflattened sinusoidal case).

Upon energizing the ballast circuit, the positive base drive circuitautomatically conducts and supplies base current to the power transistor25, thus applying line voltage to the lamp. For a 208-277 line, the peakvoltage is sufficient to start a mercury lamp. The starting lamp currentis momentarily relatively high because the reference signal is initiallyhigh due to the long RC circuit time constant of the resistor 64 and theresistor and capacitor in peak detector circuit 61. The high startingcurrent quickly heats the cathode of the lamp to avoid the undesirableglow-to-arc mode, and the current typically ramps down from a 5 ampspeak to the normal 3 amps peak in about 8 seconds. The buildup in lampcurrent is sensed by the sensing resistor 31' and supplied as anegative-going sensor signal to the negative input of comparator 35. TheRC filter 72, 73 prevents fast switching transients from giving a falsepeak lamp current signal. Assuming that the line voltage is high enoughto cause ignition of the mercury lamp (see FIGS. 4 and 6), the lampcurrent is thereafter shaped in accordance with the flattened sinusoidalreference voltage until near the end of the cycle in the valley regionof the pulsating I20 Hz d-c voltage. In the steady state, the basecurrent of the power transistor 25 is at all times proportional to thecollector current whose envelope varies approximately as half sinusoid.It will be recalled that the comparator 35 has a hysteresischaracteristic and that there is a polarity inversion since thereference signal is negative-going while the lamp current is positive.Assuming that power transistor 25 is conducting and the lamp current isincreasing, with a low output from comparator 35, the lamp currentincreases until the current sensor signal is equal to the referencesignal control band limit corresponding to maximum current (see FIG. 5).The output of comparator 35 new changes to the high output and rendersconductive the transistor 76, which is the interface between the logiccircuitry and the power transistor base drive circuit, thereby causingthe negative base drive circuit 86 to conduct while simultaneouslyturning off the positive base drive circuit 85. Power transistor 25 isnow nonconductive. and load current now circulates through the coastingpath provided by the forward biased power diode 26 and begins todecrease. in the meantime, the amount of feedback voltage from theoutput of comparator 35 to the positive input has changed, therebyswitching the basis for comparison to the other reference signal controlband limit corresponding to the minimum current value. As previouslyexplained, power transistor turns off before the end of the highfrequency cycle and is held in the nonconducting condition by the factthat transistor 76 and Darlington amplifier 91 remain conducting toapply a negative potential to the base electrode 89 of power transistor25. When the decreasing current sensor sig nal becomes equal to theother control band limit, the output of comparator 35 goes low, therebyturning off the interface transistor 76 and the negative base drivecircuit 86 while rendering conductive the positive base drive circuit85.

Primarily because the rate of rise of load current is variable since itis determined primarily by the difference between the instantaneoussinusoidal supply voltage (about 400 volts peak) and the lamp voltage(about 130 volts constant, except for the rapid increase and decrease atignition), the switching frequency of power transistor 25 automaticallyvaries from approximately lO kHz to kHz and back to 10 kHz over acomplete cycle. in a mercury lamp ballast, this sweeping of frequency,which occurs automatically and is inherent in the operation of thecircuit, helps eliminate acoustic resonance problems. As was previouslyexplained, the other pair of center-tapped secondary windings S3 and S4of transformer 45 supplies, via the second bridge rectifier 80,relatively low voltage, high current (l2 volts peak, 1 amp peak). fullwave rectified, pulsating 120 Hz unidirectional voltage to thetransistor base drive circuit. in the valley regions when the powercircuit supply voltage goes low, the comparator and associated controllogic does not function, and the local energy storage capacitor 81discharges to supply base current through the resistors 88 and 90 andDarlington amplifier 87 to the base electrode 89 of power transistor 25.This maintenance of lamp current in the valley regions is desirable forgood lamp maintenance. The positive and negative low voltage powersupply for comparator 35, provided by the series pass transistorregulators including elements 67-69 and (ST-69', does not function andthus the comparator does not function in the valley regions when thecontrol voltage provided by bridge rectifier 49 goes low, however thismakes no difference since the power voltage is low and the powertransistor is maintained in the conducting condition.

By shaping and forcing the lamp current to a flattcned sinusoidalwaveshape as shown in FIG. 4, the line current is in phase with the linevoltage and is electronically shaped to obtain a high power factorexceeding 90 percent. By properly selecting the magnitude of the reference signal according to the power level desired and by using acontrol function to obtain an electronically variable gain, lamp currentis regulated for a nominal line voltage of 277 volts to less thanone-half percent for a plus or minus 10 percent line voltage variation.The magnitude of the high frequency ripple in the lamp current (see FIG.6) is preselected and can be variable. and for this circuit hasapproximately 0.25 ampere ripple about the nominal value. The basicchopper thus can be used for mercury vapor lamps having differentwattage values by properly tailoring the resistors 88 and 92 in the basedrive circuitry, changing the value of sensing resistor 31, and byadjusting the values of the appropriate resistors in control functiongenerator 34 to change the magnitude of the reference signal accordingto the size of the lamp being powered. With this solid state ballastcircuit, lamp operation is sustained down to percent of rated linevoltage. Other advantages previously mentioned are that the choppingfrequency is automatically variable to help avoid acoustic resonanceeffects and lamp flicker.

By sustaining a minimum lamp current in the valley regions of the 120 Hzlamp current waveform, the resulting lamp voltage waveform is moresuitable for lamp reignition in each cycle and promotes prolonged lamplife. The provision of a momentarily high starting current for themercury lamp minimizes electrode degradation during arc initiation andeliminates the undesirable cathode glow-to-arc mode. The glowto-arc modeputs a high voltage and high current on the cathode. The chopper ballastoperates over a 30C to +C ambient temperature range. In this regard, andof importance to the potential commercial attractiveness of the ballast,is the fact that the high frequency circuit operation is achieved withminimum capacitive energy storage so as to eliminate electrolyticcapacitance and their associated problems. Furthermore, this circuitoperates reliably under either short circuit or open circuit lamp loadconditions. In the event that a short circuit in the lamp occurs, thecircuit operates inherently to keep the current in power transistor 25within its control limits, and in the event of an open circuit, voltageis continuously applied to the lamp terminals so that the circuitrestarts automatically, assuming that the mercury lamp is cold or hascooled down enough so that it will restart immediately.

In summary, an improved chopper ballast is particularly suitable foroperation of mercury vapor lamps from commercially available 60 Hzsingle phase line voltage in an advantageous transistor d-c chopperconfiguration that eliminates the need for bulky transformers,inductors, large correction and energy storage capacitors, undesirableelectrolytic capacitors, and power frequency filtering. in addition tothe fundamental requirement of high power factor and good regulation,the circuit supplies a lamp current waveshape especially suited formercury and other gaseous discharge lamps operated on high frequencyripple current, with provision for a good starting current waveform,automatic sweeping of the chopping frequency to eliminate acousticresonance problems, and a minimum lamp current in the valley regions ofthe pulsating energizing voltage for improved reignition. The newchopper ballast is economical, light-weight, has low volume, and can bebuilt with state-of-the-art solid state devices.

While the invention has been particularly shown and described withreference to a preferred embodiment thereof, it will be understood bythose skilled in the art that the foregoing and other changes in formand detail may be made therein without departing from the spirit andscope of the invention.

What is claimed is:

l, A solid state ballast circuit for gaseous discharge lamps comprisinga solid state chopper circuit for energization by low frequencyalternating-current line voltage and line current and including highfrequency filter means for supplying sinusoidal power voltage between apair of supply terminals and controlled switching means and coastingdevice means effectively cou pled to said supply terminals and coastinginductor means to be conductive alternately to supply lamp currentthrough said coasting inductor means to a gaseous discharge lamp,

current sensor means coupled to sense the instantaneous lamp current andproduce a sensor signal indicative thereof,

a control circuit comprising generating means for generating a referencesignal with a preselected waveshape and magnitude to determine the powerlevel and to effect shaping of said lamp current and therefore the linecurrent to obtain a high power factor, comparing means for effectivelycomparing said sensor and reference signals and producing an outputsignal, and means actuated by said output signal for supplying turn-onand turn-off signals to operate said controlled switching means at avariable high frequency chopping rate and shape said lamp current asdetermined by said reference sig nal,

said control circuit further comprising means for temporarily shapingand increasing the magnitude of said reference signal at start-up toobtain a high starting lamp current, and

said control and chopper circuits further comprising means for supplyingminimum lamp current for good reignition characteristics during the lowvoltage regions of said line voltage in each cycle when said comparingmeans is ineffective to shape the lamp current.

2. A solid state ballast according to claim 1 additionally including anadaptive control circuit connected with said generating means to furthershape said reference signal according to a selected control.

3. A solid state ballast according to claim I wherein said generatingmeans is comprised by a transformer energized by the low frequency linevoltage for deriving a control voltage, and control function generatormeans for regulating and shaping said control voltage according to apredetermined control function to produce said reference signal,

said control function generator means further including said means fortemporarily shaping said reference signal at start-up to obtain a highstarting lamp current.

4. A solid state ballast according to claim 3 wherein said comparingmeans includes a comparator with hysteresis, and said comparator has aseries pass transistor regulator low voltage power supply energized bysaid transformer which supplies regulated clipped voltage to saidcomparator except during the low voltage regions of the line voltage.

5. A solid state ballast according to claim 1 wherein said generatingmeans is comprised by means energized by the low frequency line voltagefor deriving a control voltage in phase with the line voltage, andcontrol function generator means for shaping said control voltageaccording to a predetermined control function to produce said referencesignal,

said control function generator means further including said means fortemporarily shaping said reference signal at start-up to obtain the highstarting lamp current.

6. A solid state ballast according to claim 5 wherein said means fortemporarily shaping said reference sigmil at start-up is provided by along time constant resistor-capacitor network for temporarily modifyingoperation of said control function generator means.

7. A solid state ballast according to claim 6 wherein said comparingmeans includes a comparator with hysteresis, and said comparator has lowvoltage power supply means energized by said means for deriving acontrol voltage which supplies regulated voltage to said comparatorexcept during the low voltage regions of the line voltage.

8. A solid state ballast circuit for gaseous discharge lamps comprisinga solid state chopper circuit for energization by low frequencyalternating-current line voltage and line current and including fullwave rectifying means and high frequency filter means for supplyingrectified sinusoidal power voltage between a pair of unidirectionalvoltage supply terminals, and further including a power transistor andcoasting diode connected in series between said supply terminals thatconduct alternately and supply lamp current through a coasting inductorand gaseous discharge lamp that in turn are connected in series acrosssaid coasting diode,

current sensor means coupled to sense the instantaneous lamp current andproduce an instantaneous sensor signal indicative thereof,

a control circuit comprising a transformer and a first bridge rectifierenergized by the low frequency line voltage for generating full waverectified sinusoidal control voltage, a control function generatorcircuit for shaping said control voltage and generating a symmetricallycurved reference signal with a waveshape and magnitude selected todetermine the power level and to effect shaping of the lamp current andtherefore the line current to obtain a high power factor in excess ofpercent. a comparator circuit with hysteresis for eflectively comparingsaid sensor and reference signals and producing an output signal, asecond bridge rectifier connected to said transformer for deriving fullwave rectified sinusoidal base drive power supply voltage, a positiveand negative base drive circuit connected to said second bridgerectifier for supplying alternate turn-on and turn-off signals to saidpower transistor with a base current proportional to collector current,and means for coupling said comparator output signal to energize saidnegative base drive circuit and de-energize said positive base drivecircuit to thereby operate said power transistor at a variable highfrequency chopping rate and effect shaping of the lamp current asdetermined by said reference signal waveshape,

said control circuit further comprising means for temporarily shapingand increasing the magnitude of said reference signal at start-up toobtain a high starting lamp current, and

said control and chopper circuits further comprising means for supplyingminimum lamp current for good reignition characteristics during thevalleys of the rectified sinusoidal power voltage when said comparatorcircuit is ineffective to shape the lamp current.

9. A solid state ballast according to claim 8 wherein said comparatorcircuit has a low voltage power supply circuit energized by saidtransformer that is operative to clip the full wave rectified controlvoltage at a selected low voltage level and supply power to saidcomparator circuit except during the valleys of the rectified sinusoidalpower voltage.

10. A solid state ballast according to claim 8 wherein said choppercircuit has a pair of input terminals and said high frequency filtermeans includes a shunt capacitor and series inductor connected betweensaid input terminals and full wave rectifying means, and apolycrystalline varistor connected between the input terminals of saidfull wave rectifying means, said transformer likewise having a primarywinding connected between the input terminals of said full waverectifying means, to thereby provide filtering and protection for boththe control circuit and chopper circuit.

1 l. A solid state ballast according to claim 8 wherein said comparatorcircuit has at least one series pass transistor regulator low voltagepower supply circuit energized by said transformer that is operative toclip the full wave rectified control voltage at a selected low voltagelevel and supply power to said comparator circuit except during thevalleys of the rectified sinusoidal power voltage.

12. A solid state ballast according to claim 8 wherein said controlfunction generator circuit is comprised by a resistive voltage dividercircuit having a variable resistance branch including and controlled byan insulated-gate field effect transistor, and

said means for temporarily shaping said reference signal at start-up toobtain a high starting lamp current is provided by a long time constantresistorcapacitor network connected to the gate of said field effecttransistor for temporarily determining the gate voltage and thereforethe resistance of said field effect transistor. 1

13. A solid state ballast according to claim 8 wherein said controlfunction generator circuit is comprised by a resistive voltage dividercircuit having a variable resistance branch including and controlled byan insulated-gate field effect transistor. and further comprises acontrol voltage peak detector circuit connected to the gate of saidfield effect transistor for determining the gate voltage and thereforethe resistance of said field effect transistor to thereby provideautomatic gain control of said reference signal to regulate the lampcurrent,

said peak detector circuit having a long time constant to therebyprovide said means for temporarily shaping said reference signal atstart-up to obtain a high starting lamp current.

14. A solid state ballast according to claim 8 wherein said positivebase drive circuit is connected to said second bridge rectifier to benormally conductive in the absence of said comparator output signal, and

said means for supplying minimum lamp current during the valleys of therectified sinusoidal power voltage includes local energy storagecapacitors coupled to said second bridge rectifier and to a referencepoint that discharge to provide base current to maintain conductivity ofsaid power transistor during the power voltage valleys, said highfreguency filter means including a filter capacitor connected betweensaid unidirectional voltage supply terminals which supplies lamp currentduring the power voltage valleys.

15. A solid state ballast according to claim 8 wherein said controlfunction generator circuit is comprised by a resistive voltage dividercircuit having a variable resistance branch including an insulated-gatefield effect transistor, and

an adaptive control circuit connected to the gate of said field effecttransistor to determine the gate voltage and therefore the resistance ofsaid field effect transistor to further shape said reference signalaccording to a selected control.

1. A solid state ballast circuit for gaseous discharge lamps comprising a solid state chopper circuit for energization by low frequency alternating-current line voltage anD line current and including high frequency filter means for supplying sinusoidal power voltage between a pair of supply terminals and controlled switching means and coasting device means effectively coupled to said supply terminals and coasting inductor means to be conductive alternately to supply lamp current through said coasting inductor means to a gaseous discharge lamp, current sensor means coupled to sense the instantaneous lamp current and produce a sensor signal indicative thereof, a control circuit comprising generating means for generating a reference signal with a preselected waveshape and magnitude to determine the power level and to effect shaping of said lamp current and therefore the line current to obtain a high power factor, comparing means for effectively comparing said sensor and reference signals and producing an output signal, and means actuated by said output signal for supplying turn-on and turnoff signals to operate said controlled switching means at a variable high frequency chopping rate and shape said lamp current as determined by said reference signal, said control circuit further comprising means for temporarily shaping and increasing the magnitude of said reference signal at start-up to obtain a high starting lamp current, and said control and chopper circuits further comprising means for supplying minimum lamp current for good reignition characteristics during the low voltage regions of said line voltage in each cycle when said comparing means is ineffective to shape the lamp current.
 2. A solid state ballast according to claim 1 additionally including an adaptive control circuit connected with said generating means to further shape said reference signal according to a selected control.
 3. A solid state ballast according to claim 1 wherein said generating means is comprised by a transformer energized by the low frequency line voltage for deriving a control voltage, and control function generator means for regulating and shaping said control voltage according to a predetermined control function to produce said reference signal, said control function generator means further including said means for temporarily shaping said reference signal at start-up to obtain a high starting lamp current.
 4. A solid state ballast according to claim 3 wherein said comparing means includes a comparator with hysteresis, and said comparator has a series pass transistor regulator low voltage power supply energized by said transformer which supplies regulated clipped voltage to said comparator except during the low voltage regions of the line voltage.
 5. A solid state ballast according to claim 1 wherein said generating means is comprised by means energized by the low frequency line voltage for deriving a control voltage in phase with the line voltage, and control function generator means for shaping said control voltage according to a predetermined control function to produce said reference signal, said control function generator means further including said means for temporarily shaping said reference signal at start-up to obtain the high starting lamp current.
 6. A solid state ballast according to claim 5 wherein said means for temporarily shaping said reference signal at start-up is provided by a long time constant resistor-capacitor network for temporarily modifying operation of said control function generator means.
 7. A solid state ballast according to claim 6 wherein said comparing means includes a comparator with hysteresis, and said comparator has low voltage power supply means energized by said means for deriving a control voltage which supplies regulated voltage to said comparator except during the low voltage regions of the line voltage.
 8. A solid state ballast circuit for gaseous discharge lamps comprising a solid state chopper circuit for energization by low frequency alternating-current line voltage and line current and including full wave rectifying means and high frequenCy filter means for supplying rectified sinusoidal power voltage between a pair of unidirectional voltage supply terminals, and further including a power transistor and coasting diode connected in series between said supply terminals that conduct alternately and supply lamp current through a coasting inductor and gaseous discharge lamp that in turn are connected in series across said coasting diode, current sensor means coupled to sense the instantaneous lamp current and produce an instantaneous sensor signal indicative thereof, a control circuit comprising a transformer and a first bridge rectifier energized by the low frequency line voltage for generating full wave rectified sinusoidal control voltage, a control function generator circuit for shaping said control voltage and generating a symmetrically curved reference signal with a waveshape and magnitude selected to determine the power level and to effect shaping of the lamp current and therefore the line current to obtain a high power factor in excess of 90 percent, a comparator circuit with hysteresis for effectively comparing said sensor and reference signals and producing an output signal, a second bridge rectifier connected to said transformer for deriving full wave rectified sinusoidal base drive power supply voltage, a positive and negative base drive circuit connected to said second bridge rectifier for supplying alternate turn-on and turn-off signals to said power transistor with a base current proportional to collector current, and means for coupling said comparator output signal to energize said negative base drive circuit and de-energize said positive base drive circuit to thereby operate said power transistor at a variable high frequency chopping rate and effect shaping of the lamp current as determined by said reference signal waveshape, said control circuit further comprising means for temporarily shaping and increasing the magnitude of said reference signal at start-up to obtain a high starting lamp current, and said control and chopper circuits further comprising means for supplying minimum lamp current for good reignition characteristics during the valleys of the rectified sinusoidal power voltage when said comparator circuit is ineffective to shape the lamp current.
 9. A solid state ballast according to claim 8 wherein said comparator circuit has a low voltage power supply circuit energized by said transformer that is operative to clip the full wave rectified control voltage at a selected low voltage level and supply power to said comparator circuit except during the valleys of the rectified sinusoidal power voltage.
 10. A solid state ballast according to claim 8 wherein said chopper circuit has a pair of input terminals and said high frequency filter means includes a shunt capacitor and series inductor connected between said input terminals and full wave rectifying means, and a polycrystalline varistor connected between the input terminals of said full wave rectifying means, said transformer likewise having a primary winding connected between the input terminals of said full wave rectifying means, to thereby provide filtering and protection for both the control circuit and chopper circuit.
 11. A solid state ballast according to claim 8 wherein said comparator circuit has at least one series pass transistor regulator low voltage power supply circuit energized by said transformer that is operative to clip the full wave rectified control voltage at a selected low voltage level and supply power to said comparator circuit except during the valleys of the rectified sinusoidal power voltage.
 12. A solid state ballast according to claim 8 wherein said control function generator circuit is comprised by a resistive voltage divider circuit having a variable resistance branch including and controlled by an insulated-gate field effect transistor, and said means for temporarily shaping said reference signal at start-up to obtain a high starting lamp current is providEd by a long time constant resistor-capacitor network connected to the gate of said field effect transistor for temporarily determining the gate voltage and therefore the resistance of said field effect transistor. l
 13. A solid state ballast according to claim 8 wherein said control function generator circuit is comprised by a resistive voltage divider circuit having a variable resistance branch including and controlled by an insulated-gate field effect transistor, and further comprises a control voltage peak detector circuit connected to the gate of said field effect transistor for determining the gate voltage and therefore the resistance of said field effect transistor to thereby provide automatic gain control of said reference signal to regulate the lamp current, said peak detector circuit having a long time constant to thereby provide said means for temporarily shaping said reference signal at start-up to obtain a high starting lamp current.
 14. A solid state ballast according to claim 8 wherein said positive base drive circuit is connected to said second bridge rectifier to be normally conductive in the absence of said comparator output signal, and said means for supplying minimum lamp current during the valleys of the rectified sinusoidal power voltage includes local energy storage capacitors coupled to said second bridge rectifier and to a reference point that discharge to provide base current to maintain conductivity of said power transistor during the power voltage valleys, said high freguency filter means including a filter capacitor connected between said unidirectional voltage supply terminals which supplies lamp current during the power voltage valleys.
 15. A solid state ballast according to claim 8 wherein said control function generator circuit is comprised by a resistive voltage divider circuit having a variable resistance branch including an insulated-gate field effect transistor, and an adaptive control circuit connected to the gate of said field effect transistor to determine the gate voltage and therefore the resistance of said field effect transistor to further shape said reference signal according to a selected control. 